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參數資料
| 型號： | MC34065P-L |
| 廠商： | MOTOROLA INC |
| 元件分類： | 穩壓器 |
| 英文描述： | HIGH PERFORMANCE DUAL CHANNEL CURRENT MODE CONTROLLERS |
| 中文描述： | 0.4 A DUAL SWITCHING CONTROLLER, 500 kHz SWITCHING FREQ-MAX, PDIP16 |
| 封裝： | PLASTIC, DIP-16 |
| 文件頁數： | 6/16頁 |
| 文件大小： | 419K |
| 代理商： | MC34065P-L |
MC34065–H, L MC33065–H, L
OPERATING DESCRIPTION
6
MOTOROLA ANALOG IC DEVICE DATA
The MC34065–H,L series are high performance, fixed
frequency, dual channel current mode controllers specifically
designed for Off–Line and dc–to–dc converter applications.
These devices offer the designer a cost effective solution with
minimal external components where independent regulation
of two power converters is required. The Representative
Block Diagram is shown in Figure 15. Each channel contains
a high gain error amplifier, current sensing comparator, pulse
width modulator latch, and totem pole output driver. The
oscillator, reference regulator, and undervoltage lock–out
circuits are common to both channels.
Oscillator
The unique oscillator configuration employed features
precise frequency and duty cycle control. The frequency is
programmed by the values selected for the timing
components RT and CT. Capacitor CT is charged and
discharged by an equal magnitude internal current source
and sink, generating a symmetrical 50 percent duty cycle
waveform at Pin 2. The oscillator peak and valley thresholds
are 3.5 V and 1.6 V respectively. The source/sink current
magnitude is controlled by resistor RT. For proper operation
over temperature it must be in the range of 4.0 k
to 16 k
as
shown in Figure 1.
As CT charges and discharges, an internal blanking pulse
is generated that alternately drives the center inputs of the
upper and lower NOR gates high. This, in conjunction with a
precise amount of delay time introduced into each channel,
produces well defined non–overlapping output duty cycles.
Output 2 is enabled while CT is charging, and Output 1 is
enabled during the discharge. Figure 2 shows the Maximum
Output Duty Cycle versus Oscillator Frequency. Note that
even at 500 kHz, each output is capable of approximately
44% on–time, making this controller suitable for high
frequency power conversion applications.
In many noise sensitive applications it may be desirable to
frequency–lock the converter to an external system clock.
This can be accomplished by applying a clock signal as
shown in Figure 17. For reliable locking, the free–running
oscillator frequency should be set about 10% less than the
clock frequency. Referring to the timing diagram shown in
Figure 16, the rising edge of the clock signal applied to the
Sync input, terminates charging of CT and Drive Output 2
conduction. By tailoring the clock waveform symmetry,
accurate duty cycle clamping of either output can be
achieved. A circuit method for this, and multi–unit
synchronization, is shown in Figure 18.
Error Amplifier
Each channel contains a fully–compensated Error
Amplifier with access to the inverting input and output. The
amplifier features a typical dc voltage gain of 100 dB, and a
unity gain bandwidth of 1.0 MHz with 71
°
of phase margin
(Figure 5). The noninverting input is internally biased at 2.5 V
and is not pinned out. The converter output voltage is
typically divided down and monitored by the inverting input
through a resistor divider. The maximum input bias current is
–1.0
μ
A which will cause an output voltage error that is equal
to the product of the input bias current and the equivalent
input divider source resistance.
The Error Amp output (Pin 5, 12) is provided for external
loop compensation. The output voltage is offset by two diode
drops (
≈
1.4 V) and divided by three before it connects to the
inverting input of the Current Sense Comparator. This
guarantees that no pulses appear at the Drive Output (Pin 7,
10) when the error amplifier output is at its lowest state (VOL).
This occurs when the power supply is operating and the load
is removed, or at the beginning of a soft–start interval
(Figures 20, 21).
The minimum allowable Error Amp feedback resistance is
limited by the amplifier’s source current (0.5 mA) and the
output voltage (VOH) required to reach the comparator’s 1.0 V
clamp level with the inverting input at ground. This condition
happens during initial system startup or when the sensed
output is shorted:
Rf(min)
≈
3.0 (1.0 V)
0.5 mA
1.4 V
= 8800
Current Sense Comparator and PWM Latch
The MC34065 operates as a current mode controller,
whereby output switch conduction is initiated by the oscillator
and terminated when the peak inductor current reaches the
threshold level established by the Error Amplifier output.
Thus the error signal controls the peak inductor current on a
cycle–by–cycle basis. The Current Sense Comparator–PWM
Latch configuration used ensures that only a single pulse
appears at the Drive Output during any given oscillator cycle.
The inductor current is converted to a voltage by inserting a
ground–referenced sense resistor RS in series with the
source of output switch Q1. This voltage is monitored by the
Current Sense Input (Pin 6, 11) and compared to a level
derived from the Error Amp output. The peak inductor current
under normal operating conditions is controlled by the
voltage at Pin 5, 12 where:
V(Pin 5, 12) – 1.4 V
Ipk =
3 RS
Abnormal operating conditions occur when the power
supply output is overloaded or if output voltage sensing is
lost. Under these conditions, the Current Sense Comparator
threshold will be internally clamped to 1.0 V. Therefore the
maximum peak switch current is:
Ipk(max) =
1.0 V
RS
When designing a high power switching regulator it may
be desirable to reduce the internal clamp voltage in order to
keep the power dissipation of RS to a reasonable level. A
simple method to adjust this voltage is shown in Figure 19.
The two external diodes are used to compensate the internal
diodes, yielding a constant clamp voltage over temperature.
Erratic operation due to noise pickup can result if there is an
excessive reduction of the Ipk(max) clamp voltage.
A narrow spike on the leading edge of the current
waveform can usually be observed and may cause the power
supply to exhibit an instability when the output is lightly
loaded. This spike is due to the power transformer
interwinding capacitance and output rectifier recovery time.
The addition of an RC filter on the Current Sense input with a
time constant that approximates the spike duration will
usually eliminate the instability, refer to Figure 24.
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